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Index |
aspects of wide-band,
linear HF power amplifiers Introduction When designing wide-band, linear HF power
amplifiers a small number of basic rules should be considered. These are not
very complicated but should be followed rather carefully when the object is
to maximize power and efficiency. With these basic rules firmly into
position, further details will follow quite naturally. These basic design rules apply to LF and HF
amplifiers although each of these fields also exhibit additional requirements
that are specific to the particular range of frequencies. With basic design
rules comparable, still many radio-amateurs have no problems designing LF
amplifiers and will shy away from their home-brow linear HF power
equivalents. The short notes in this article will discuss these basic design
rules for radio-hams to start thinking about designing their own amplifiers
from available components, probably already present in the junk box. In tuned, single frequency amplifiers the
choice of the system impedance is less important as most internal parasitic
impedances will be part of a designed-in impedance that will be tuned to the
desired operating frequency. In wide-band systems, system impedance Z0
is one of the primary selections. This impedance in principle is a
freely selectable quantity but somewhat restrained in practice. As a general rule, the higher Z0,
the lower system currents and therefore the lower the loss in all parasitic
series inductances, resistances and further 'additional', series impedances.
This premium is lost however in all parasitic (parallel) capacitances that
will be unavoidable and will limit maximum frequency. At a low system impedance, system currents
will be high(er) and therefore all series-loss will
increase. Because of the low impedance, parallel capacitance will be less
important allowing for higher maximum frequencies. From the above it follows system impedance
selection to be connected to the desired frequency and frequency range of the
amplifier, with the lower impedance range related to the higher operational
frequencies. Figure 1 is presenting a general idea about the position of the
system impedance, that does not have to be represented by an actual component
and in practice usually isn't either.
System impedance also is an entity of choice
and therefore be closely related to practical considerations e.g. availability
and price of components designed for a particular characteristic impedance.
To this extend one should think of transmission-lines, filters, sub-systems
etc., but also that high-impedance inductors may be constructed from lower
diameter, un-silvered wire and low-voltage capacitors usually also mean lower
price. Although a system impedance of 50 Ohm for wide-band systems is a well
known 'standard', 75 Ohm is a very good option and other impedances may be
even better depending on local requirements. At wide-band power amplifiers based on vacuum
tubes it may be profitable to select a higher system impedance. A vacuum tube
usually operates under a high-voltage, low(er)
current 'regime' making matching impedances also of a higher level. Further more,
(dipole) antennas only exhibit a rather low termination impedance (around 50
Ohm) when resonating and a much higher impedance outside resonance. Selecting
a high 'system impedance' when matching an antenna to a tube amplifier
therefore usually will require a lower transformer ratio which in general is
more profitable from an efficiency point of view and also simpler to
construct. At wide-band power amplifiers based on
semi-conductors a low impedance system impedance may be more profitable since
transistors usually operate in a low-voltage high current regime. To match to
a high(er) impedance antenna, an in-between step to
an 'intermediate' system impedance usually is a practical solution to avoid
high-step-up ratio's. At power amplifiers a wide range of
voltage-current regimes (impedance-level) may be encountered, amongst
others depending of the type of active
component. From a system point of view it is efficient to arrive at system-impedance
level as soon as this is practical and to this extend a matching transformer
usually is the component of choice. Parasitic effects are always a point of
concern in wide band amplifiers. Therefore impedance matching ratio's should
not be too high to avoid parasitic effects of the low impedance regime to
influence the circuit at the high-impedance site and vise versa. In this
respect one could think of (serial, leak) inductance as a factor of concern
in the low-impedance regime and parallel and inter-winding capacitive effects
at the higher impedance site. In practice and for high(er
frequency wide-band amplifiers in particular, impedance ratio's of 1 : 16
already are very high, while 1 : 25 for LF amplifiers are run of the mill and
very low when using vacuum tubes. What parallel
impedance? A matching transformer is to match the lower
impedance regime to the higher environment, preferably without being a
'matching factor' in itself. The transformer therefore should be more or less
'invisible' at system level, meaning to introduce as little loss and/or phase
shift as possible. A practical rule of thumb to accomplish this is the
transformer to exhibit four times the system impedance at the lowest
operational frequency: Zt = 4 x Z0, At this parallel impedance, standing-wave ratio
will deteriorate to SWR 1,28 but will improve at higher frequencies to become
even more 'invisible'. The number of turns to arrive at this impedance
depends on the type of transformer core material, for which the manufacturer
will supply winding factor Note! Most manufacturers define
winding factor TN36/23/15-4C65 toroide ( T157-2
toroide (1.57" ( Recalculating the latter to the basic
definition will yield: How many turns? To calculate the number of turns for a
specific application, we may rework the equation: Zt
= w . n = 0,8 √ ( Z0 / ( f . with: n =
number of turns (at system impedance level) Z0
= system impedance f =
lower operating frequency (Hertz) Selecting 50 Ohm as the system impedance and
a n = 13,7 √(1/f) (f in MHz. = lower operating frequency) which would lead to 10 turns for a lower
operating frequency of 2 MHz. Note. More turns is not necessarily
the better. At more turns than the required number the parasitic parallel
capacitance will increase lowering the maximum operating frequency. What power? Depending on the actual operating frequency,
the transformer core will be more or less lossy. Depending on the voltage
across the transformer, some power will be dissipated in this core-loss and
will heat-up the core. Several mechanisms will limit the maximum core
temperature. Maximum temperature * Curie temperature. This number is related
to the maximum ferrite temperature, above which the material will lose all
permeability. This implies impedance to drop to very low values which usually
will destroy the component and / or the system it was embedded in. When the
system is going first, temperature will drop and material will return to the
usual permeability. This Curie temperature is very much related to the
specific ferrite material, e.g. * Mechanical degradation. This is related to
the maximum core temperature of non-sintered materials e.g. electrolytic-iron
(hydrogen reduced) iron powder and Carbonyl iron powder. According to some manufacturers,
these materials will be permanently degraded when exposed to temperatures
above * Winding materials temperature. Some
transformers will be constructed using transmission-lines, as in
transmission-line transformers. These lines are usually constructed from a
soft, easily deformable material that will loose mechanical integrity above In general maximum temperature rise from
internal heating for transformers and baluns is restricted to 30 K to also
allow for high environmental temperature conditions as found inside power
amplifier cabinets and at the antenna at a hot summer day. Maximum voltage A transformer at a In the example transformer above, that
required 10 turns for reasons of system impedance, maximum allowable voltage
is: 10 x 42 Volt = 420 Volt at 2
MHz., leading to
over 3,5 kW of system power in
a 50 Ohm system or maximum 10 x 24 Volt = 240 Volt at 30 MHz., leading to
around1,15 kW. When constructed from RG58 coaxial transmission-line, it will
be this latter that limits maximum system power to 350 W. at 30 MHz. for
reasons of maximum current handling capacity. We prefer to discuss maximum voltage across
the transformer over specifying maximum allowable system power. A well
designed transformer may still be destroyed at much lower system power when
system impedance is higher than designed for, as in a dipole antenna outside
resonance. Within power limits, maximum voltage across the transformer may
become easily too high leading to much higher internal core dissipation and
finally destruction of this component and even the system driving this. In table 1 an impression may be obtained of
the maximum allowable voltage across a one-turn inductor at the well known Z = n2 x Zt. Maximum voltage for more turns may be
obtained by multiplying the voltage of this one turn inductor by the number
of turns as in: V = n x Vm.diss.
Maximum allowable system power may be obtained
by taking the maximum, more turns voltage squared and dividing by the system
impedance as in: Ptrafo max. =
(n x Vm.diss)2 / Z0.
The second column is the ferri-magnetic
resonance frequency (fr) of the specific
ferrite material. This is a materials parameter and is related to the reactance
(μ') and loss (μ")
parameters as the frequency where: μ' = μ" (Q = 1). It is
inversely related to permeability, so 3E25 type of material representing the
highest permeability and is therefore a low frequency material. From table1 it may be seen two ferrite
parameters to compete when determining maximum voltage across the inductor,
i.e. permeability and loss, that both are frequency dependent, but in
different ways. When finally loss will be the determining factor at a
frequency above fr, maximum allowable
voltages tend to arrive in the same ball-park. Also, total impedance will still continue to
rise long after this ferri-magnetic resonance frequency, which is very useful
when designing choke type of applications. The number of turns at the 'non-system side'
of the transformer, and so the transformer ratio 'T', is still un-determined,
but will follow from the system to match to. We will attend to this in the
next paragraph. Independent of T, most important transformer parameter
already have been determined like the core material, number of 'system-side'
turns and maximum allowable voltage (system power). The transformer will match one impedance to
an other, but will not influence the power ratio from input to output, apart
from a small amount of power loss. This is a 'no-brainer' but will often be
forgotten when designing transformers. We will use this simple 'input power
is equal to output power' rule quite often. Background to various ferrite calculations
may be found in the series of articles on this subject as in: Ferrites in HF
applications' on this web-site. What transformer
type? We discussed various transformer aspects, that
may take different shapes depending on materials and shapes available. Also
we discussed 'turns' and transformer ratio's in terms of the classic flux
transformer, i.e. transformers where input is coupled to the output by means
of the electro-magnetic field in the core. This transformer type will serve
its purpose if well designed but is also suffering from inherent limitations,
related to the mechanism of flux coupling and parasitic capacitance across
and between 'windings'. Flux transformers When core loss is becoming noticeable as a
system parameter, i.e. around and after ferri-magnetic resonance of the core
material, inter winding coupling is going down. This will be noticeable in
the increasing of leakage flux, translating into a series inductance with the
transformer which will limit high(er) frequency
response. Together with the intrinsic transformer loss, this type of
transformers will find their application limit around the ferri-magnetic
resonance frequency, i.e. when μ' = μ" or Q = 1. More on this
subject may be found at HF
transformers. Transmission-line
transformers A different type of transformers is
consisting of one or more transmission-lines, that are combined in series or
in parallel at the input and / or at the output. Transformer ratio is
determined by directly adding currents and / or voltages and different
transformer ratios may be created although somewhat less flexible than with
flux transformers. Because of the inherent wide-band characteristics of these
transmission-lines, this type of transformer may be applied over a much wider
range of frequencies than the flux type. Critical point at transmission-line
transformers is the input to output separation. For this application, again
core materials will be used to create a high input to output impedance. It is
virtually unimportant how this high impedance is created, by loss, by
reactance or a combination, which is leading to a much wider frequency
application range of the same core material, as also may be appreciated from
table 1. As far as 'parallel impedance' at the
transformer input is concerned, the same formula's apply as with the flux
transformers. Because of a different circuit lay-out, a different strategy
has to be followed for feeding DC-power to the active components. More on transmission-line transformers may be
found in: Transmission-line
transformers. As discussed before, power amplifiers may be
designed with vacuum-tubes or solid-state devices or even a combination of
the two. At the higher power range vacuum-tubes will usually be the preferred
component as it is less difficult to remove the internally dissipated power
that results from a less than 100 % system efficiency. Modern higher power
broadcast stations may currently be equipped with solid-state output stages,
although these have been design as a combination of a number (tens) of lower power solid-state amplifiers
'building bricks', combined to deliver the required high power output. This
will also ensure higher system reliability when only one of these building
bricks is failing. Whatever the active component, the design procedures are
very much comparable. As an example, we will start designing a
power amplifier based on a real life transistor, BUZ308, although this type
will not likely be applied in a final HF wide band-width power amplifier. The
design procedure however will not be different. Factory information The transistor manufacturer will supply a
great number of parameters and specifications. To start-off we will look at
some limiting values we should obey to exploit the life expectance for the
component. For BUZ308 we find: - maximum drain - source voltage: 800 Volt (Vmax), - maximum drain current: 2,6 Ampere (Imax)
- maximum dissipation internally: 75 Watt (Pmax). Other important information may be supplied
on the internal thermal conditions. According to specifications, maximum silicon
temperature is Tj = Next we will look at transistor
characteristics as in figure 2.
To our surprise we find numbers (far) above
the maximum figures for this component (current) or not far enough (voltage).
Usually this is meaning the component is (also) being designed for a number
of pulse-type applications or specific high-voltage low-current circuits as
in inductive switching. Further more this manufacturer has been so kind to also supply the maximum internal power
rating as in the dotted line. Knee voltage In figure 2 we find a number of horizontal
lines for various gate voltages. When we follow the line for Vg = 4,5 V from
high to low drain voltage, we find the curve to deviate from the horizontal
line at around 15 V., curving back to 0V thereafter. This 15 V. point is
called 'knee voltage' and is marking an area we should stay out for linear
applications. Calculation values Whatever choices we like to make at designing
this transistor power amplifier, we should take care to keep some margin to
allow for adverse operating conditions e.g. switching to full power at less
than ideal matching conditions (SWR > 1). Therefore we prefer to stay away
by at least a factor of 1,5 from the earlier mentioned absolute limiting
values. Our limiting values for the amplifiers we are designing therefore
will be: Id max. = Imax / 1,5 =
1,7 Ampere, Vd max. = Vmax / 1,5 = 530 Volt, Pmax
= 75 / 1,5 = 50 Watt. Class A amplifier Different types of amplifiers may be designed
around the selected transistor. We may remember class A amplifiers to operate
at low distortion, so this might be our first try. In a class A amplifier,
the active element is adjusted at the control port (at the BUZ308, the gate)
so half the maximum current is flowing in the output terminal. Load line When selecting the maximum (calculation)
drain current: Ip van 1,7 Ampere, at the
knee voltage: 20 Volt, we may draw a straight line to the point where drain
voltage is 80 Volt at drain current is 0 Ampere. This line is always below
the dotted line for maximum allowed power so may be regarded as an allowed
and safe load line. In figure 2 this is the red line. With this load line, the power amplifier is
completely determined. This may be appreciated since: - At Id = 0 Ampere, no voltage may
be found across an impedance in the drain circuit, so this voltage may be
regarded as the supply voltage: Vb = 80 Volt. - Maximum output voltage as delivered to the
load is equal to the supply voltage minus the knee voltage: Vp = Vb - Vkn = 80 - 20 Volt = 60 Volt. This is also the top-top value of the output voltage
to the load, with an effective (rms) value of: Vout = 60 / 2√2 V. = 21,2 Volt
rms. - Maximum current through the transistor (and
the load) is 1,7 Ampere and minimum current is 0 Ampere. This is the top-top
value of the output current, with an effective (rms)
value of: Iout = 1,7 / 2√2 A. = 0,6 Ampere rms. - Out of the rms
values for voltage and current we may calculate maximum power as delivered to
the load: Pout = Vout x
Iout = 21,2 x 0,6 = 12,7 Watt - Class A amplifier at no drive is set in the
middle of the load-line, so at a drain current of: IDC = Ip
/ 2 = 1,7 / 2 = 0,85 Ampere which leads at the load line to a drain
voltage of 50 V. - Total delivered DC input power to the
transistor is: PT = VDC x IDC
= 50 x 0,85 = 42,5 Watt. - The efficiency (ή) of the power amplifier may be
calculated in a percentage of the DC input ή = (Pout / PT
) x100 % = (12,7 / 42,5) x 100 % = 29,9 % This is considerably less that what we expect
from a class A amplifier (50%) and is mainly due to the relatively high
knee-voltage as compared to the supply voltage. The area below the knee is
un-usable (outside linear voltage - current behavior) but is contributing to
the total DC input power. - The load-line may be directly translated
into a load resistance, which may be calculated from AC peak voltage and
current: Rb = Vp
/ Ip = 60 / 1,7 Ohm = 35,5 Ohm, This in turn is determining the output
transformer, that will have an impedance
transformer ratio of 35,5 : 50 = 1,4 : 1 , of which the turns ratio is following
from the square route: 1,2 : 1. With 10 turns at the primary side as
calculated before, the secondary number is 10 / 1,2 = 8 turns. Thermal requirements
of the class A amplifier The thermal requirements of a power amplifier
are an important part of the design. From the manufacturers specifications we
learn maximum internal temperature: Tj = Difference between the amplifier input and
output power (see also efficiency) is turned into heat, we may calculate: PW = PT - Pout
= 42,5 - 12,7 Watt = 29,8 Watt Because of this power, a temperature gradient
will exist across the thermal resistance of: ΔT = PW x Rth
j-mb = 29,8 x 1,67 = 49,8 K Since maximum internal temperature is Tmb max = 150 - When we would like to operate this amplifier
at a maximum ambient temperature of: Tamb = Rth heatsink =
(Tmb max - Tamb) / PW = (100,2 - 35) / 29,8
= 2,12 K/W, and this is more than a
simple cooling flap. Some conclusion on
class A amplifier By selecting a class A amplifier, we obtained
a solution with a relatively low output power and a low power efficiency.
Furthermore the power supply will have to supply a relatively high average
current of It is clear the class A choice to not be an
optimal one, so we better investigate a different set-point (amplifier
class). Load line In a class B amplifier, the active element is
adjusted at the control port (at the BUZ308, the gate) so (just) no current
will flow in the output terminal. When increasing the drive, drain current
will flow in the positive half of the drive cycle and no current during the
negative half. Since our amplifier is to be operated in the linear mode, we
need two active elements that will each operate at the other half of the
input drive. Both halves will be combined in the output transformer to
generate the full output cycle. This also means each active element to
deliver full power at half the time, or on average, half power during the
full period. In the off half cycle, the active element will not dissipate any
(DC or AC) power. We may again operate each active element
according to the red load-line of figure 2, since this has been selected for
maximum drive and power as related to the absolute maximum ratings. Note the transistor this time should be rated
for double the drain voltage since the output voltage of the active device is
added to the drain voltage of the non-active device, that is at supply
voltage level at no drive. At the selection of the load line, the
complete amplifier is determined. - At Id = 0 Ampere, supply
voltage: Vb = 80 Volt. - Maximum output voltage swing: Vp = Vb - Vkn = 60 Volt. This is the peak value of one drive period,
the other half is generated by the second transistor. Effective voltage of
this half cycle is: Vout =
60 / √2 Volt = 42,4 Vrms. - Maximum current is again 1,7 Ampere peak
value, with an effective value of Iout = 1,7 / √2 Ampere = - During the half period cycle each
transistor is generating Pout
= Vout x Iout
= 42,4 x 1,2 = 50,9 Watt, and no power during the next half period when the
other transistor takes over. Total output power for this class B amplifier
therefore is also 50,9 Watt. - Through each transistor a current is
flowing only during the active cycle. Average value of this current is: IDC = 2 x Imax / p = 1,08 Ampere. DC voltage at the transistor is equal to the
supply voltage, so total DC input power per transistor, per half cycle is: P = IDC x VDC = 1,08 x
80 = 86,4 Watt, which is also total DC input power for the full drive cycle
of this class B amplifier. - Total efficiency of this class B amplifier
again is calculated as a percentage of total DC power as delivered to the
amplifier: ή = (Pout / PT) x 100 % = (50,9 /
86,4) x 100 = 58,9 %. This again is lower than the theoretical
maximum for this class of amplifiers (68%) again because of the relatively
high knee voltage. - Drain load (load-line) per transistor is
the same as in the class A amplifier, making also turns ratio (per transistor)
identical. The output transformer therefore will have a turns ratio of (1+1)
: 1,2 or (8 + 8) : Thermal requirements
of the class B amplifier Calculations are running in parallel to those
in the class A amplifier. Power as delivered to the amplifier that is not
delivered to the output, will be generated in heat and should be drained away
to below permissible temperatures. For this class B amplifier and per
transistor: PW = PT - Pout
= 43,2 - 25,5 Watt = 17,8 Watt. This will generate a temperature gradient in
the transistor from junction to mounting base: ΔT = PW x Rth
j-mb = 17,8 x 1,67 = 29,6 K The maximum allowable mounting base
temperature therefore will be Tmb max = 150 - When we would like to operate this amplifier
at a maximum ambient temperature of: Tamb = Rth heatsink =
(Tmb max - Tamb) / PW = (120,4 - 35) / 17,8
K/W = 4,8 K/W. When we prefer one mounting base for the two
transistors, thermal resistance will be halved to 2,4 K/W. Comparing class A to
class B amplifier. To compare the class A to the class B
amplifier, table 2 has been generated:
In table 2 it is immediately clear what the
efficiency difference between the two amplifier classes is bringing. At about
double the input power by the power supply, the class B amplifier is
delivering four times the output power and will require about the same
heatsink as the class A amplifier. Since each transistor in this class B
amplifier is taking DC current per cycle, no net DC current will flow though
the output transformer so full drive capabilities are available for HF-power
with inherent lower distortion. This example is showing it pays to go for
amplifiers with high efficiency because of the positive effects at various
system components. Selecting a low 'knee-voltage' transistor will further
improve efficiency. Note, all calculations have been performed at
full output power. At lower (average) HF output power as in SSB operation,
average efficiency will also be lower. Rest current In a practical class B amplifier each
transistor usually will not be adjusted to a no-current at no-drive
situation. Starting from zero drive, the first part of the transistor
characteristic is not very linear, generating harmonic content in the output
current. By adjusting each transistor to a small 'resting current' this
non-linear drive area may be avoided. This amplifier-setting is called a
class AB amplifier since the set-point is somewhere in-between class A and
class B. At this set-point however, transistors are operating in class B mode
for most of the time. The class AB setting is requiring some
additional DC power that will not be delivered to the output and so will make
total efficiency of this class somewhat lower then full class B amplifier,
with additional (small) requirements to the heatsink. Feedback A second practical point concerns the
transistor un-equality. Even from the same manufacturing batch, transistors
will always exhibit small differences as to their characteristics. This may
result in one transistor to still be completely switched-off with the other
transistor already fully at the class AB drain current for the same
set-point. A second effect may be noticeable at the output with one
transistor delivering full power and the other just keeping a 'toe in the
water'. These undesired situations we of course very much like to avoid. An effective way of dealing with these
effects is to apply some form of feed-back, for instance by having a small
portion of total drain current to deliver a voltage across a source resistor.
This voltage will effectively be in series with the input drive and
set-point, but in the opposite phase. This feed-back voltage will have a
positive and equalizing effect to transistor drive characteristics to the DC
(set-point) as well as to AC (linearity). As a rule of thumb, source voltage
should be in the range of 10 - 30 % of the gate voltage at full drive swing,
this to be applied even when transistors are selected for pairing. With these feed-back resistors, linearity
will be improved and so output harmonic content will be lowered. Since this a an HF amplifier, these source
resistances should be of a 'no-inductance' type and also of sufficient power
handling capacity since source currents may be high. Paralleling
transistors To enhance current drive capabilities, more
transistors may be connected in parallel. To ensure good tracking of
characteristics, the same principles of feed-back apply. As for the calculations (load-line, power
delivery and handling, heat considerations), the same principles apply. Because
current are doubling (for two transistors in parallel) the same
characteristics of figure 2 apply, this time with doubled current figures on
the vertical axes. With three transistors in parallel the figures obviously
triple, etc. Note, the matching transformer also to match
to a lower impedance, with half the primary impedance for two transistors in
parallel one third at three transistors etc. All further calculations will
follow the, by now familiar patterns as above. Some conclusions - With the above method it should be not too
difficult to design a power amplifier with components already available from
the junk-box (transistors, ferrite cores, heatsink etc.). - Although high linearity is always a design
requirement for wide-band amplifiers, this should be balanced to other
practical requirements as system efficiency and available drive and DC input
power. - High
system efficiency usually translates into simpler requirements to the
heatsink, power supply and will result in more output power. - The higher the amplifier class (A, AB, B,
C, currently also D and E are practical), the higher system efficiency
although not all classes may be applied in wide-band linear amplifiers. Closing remarks The next project phase is concerned with practical
system construction. To this phase, practical experience is an important
requirement and usually means battling with al sorts of parasitic and
non-ideal effects, that are becoming more important as frequency rises.
Therefore it is important for a starting designer to study the designs of
experienced people, with the principles as explained in this article as a
basis. In this way the practical learning curve will be less steep and my
help starting your own amplifier design
Bob J. van Donselaar. on9cvd@veron.nl |
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