Index
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Ferrites in HF-applications
inductors and transformers (published in Electron #11, 2001) General This chapter is the third in a series of articles
on ferrite materials in HF-applications. The first article is an introduction
to this field with an overview
of some widely applied materials and most important properties. The
second article is on materials back-ground and most important
HF-application formulas. It is advisable to read the articles in the
above order especially since each next chapter is building on information and
formula's already explained earlier and referencing to this. Ferrites in HF-inductors
In a first approximation inductors in
electronic circuits are considered as purely inductive components. Next it
appears these inductors are far from ideal and losses have to be taken into
account, in general to be expressed as a ratio of reactance to (copper) loss
in the quality factor: Q, as in formula 15. Q = XL / rs
= w.L / rs. When applying (ferrite) core materials, most
losses stem from the core material, especially when applied at HF
frequencies. In formula 16 we showed the inductor quality could be expressed
as a ratio of the inductance factor, m and core-loss factor, m of ferrite materials as specified by the manufacturer. Q
= m / m. with materials data as in table 2. Antenna-trap
inductors Core properties will influence the type of
application. HF- amateur and author, Moxon writes
in his book: "HF antenna's for all locations", `...we should be
careful at applying core materials in antenna trap inductors as these should
exhibit a quality factor of over 200, which makes ferrites in general unfit
for these applications...`. With the 'tools' from chapter 2, we may
examine Moxon's remarks. As an example we will
investigate a few properties of the antenna trap inductor of 5,4 mH as being applied in Multiband trap antenna. In this
design an inductor without core materials has been applied, for reasons that
will become obvious later on. We may consider to design a very compact trap
by applying a small 23/14/7 toroide of 4C65 material exhibiting a core factor
L
= n2 . AL
(nH). out of
which: n = 7,87 , we take n = 8 (only integer values). When calculating the antenna with EZNEC, we
find largest voltage across the trap to appear around 7 MHz, at an antenna
input power of 100 Watt to rise to 450 volt. Induction As we may find in table 2 and formula 16,
4C65 material is exhibiting a high quality factor of 125 at 7 MHz. To avoid non-linear behavior, voltage across an
inductor on ferrite material should be limited. Using formula 5 we calculate
maximum voltage across this inductor. UL (induction) =
0,89 . Bsat . f . n .
A With Bsat
= 380 mT for 4C65, n = 8 turns, f = 7 MHz. and the area for this core size A = 31 mm2,
we find a maximum voltage: UL (induction) = 590 volt. Since maximum voltage for
100 Watt is 450 volt, no non-linear behavior will occur up to this input
power. Permeability A different problem however is popping up. At
7 MHz. inductor reactance is XL = wL = 248 W. With 450 volt across, effective
current is H = n . I / l With l = Manufacturers specification (data
books) is showing that permeability m is diminishing form
125 to 70 under this magnetic field strength. Since we started our
calculations from m = 125 ( Power
dissipation
Because permeability is down to
70 with loss unchanged, quality factor for this material has changed as well: Q = m / m = 70/1 = 70. Parallel loss resistance may be calculated
from: RFp = Q . XL
= 70 . 248 = 17360 W. Since we found 450 volt across
the inductor, this amounts to 4502 / 17360 = 11,7 watt per trap,
so 23,4 Watt in total. Not only will this amount of power be lost to our
transmission, but it will probably also destroy the toroide since maximum
internal dissipation for this core volume is 4 Watt for a maximum temperature
rise of 30 K. The origin of the Moxon remark as above is now complete clear. Conclusion
In general ferrite materials are
not recommended for very precise HF tuning inductors in general (permeability
vs temperature characteristic) and in power tuning
applications in particular (internal loss). Ferrites in HF chokes
Why ferrites The impedance between inductor terminals is
the most important parameter in choke application, much more so than the
precise phase relation between current and voltage; as long as this is
impedance is high (usually at low DC resistance), the application will
function well. As an example we perform some calculations
around the 10 mH inductor we designed on a 3E25
ferrite toroide in the first chapter. In this design we started from table 1,
where we found winding factor AL = 7390 nH/n2 (based on μi
= 6000) for this material and size. Since the winding factor has been
determined at very low frequency and no loss, we have to recalculate for this
choke application and for other operating frequencies e.g. 4 MHz., using formula 14. AZ(4 MHz) = AL . mC(4 MHz) / mi Table 2 is presenting permeability (μ')
and loss factor (μ") for a number of well known
ferrite materials, including the complex assembly (μc
). Recalculating the winding factor for our choke at an operating
frequency of 4 MHz., we start from the complex
permeability factor, μc , as in
table 2 (μc = 601), so AZ(4 MHz) = AL(DC) . 601 / 6000 = 0,1 . AL (DC) (nH) Impedance at our operating frequency of 4 MHz. will follow from: |Zt| = w . nē . (0,1 . AL (DC)) At still higher frequencies, the complex
permeability, μc will be still
lower, at 30 MHz. only 90. So between 4 and 30 MHz. (factor of This is clearly demonstrating the importance
of (even relatively low frequency) ferrite materials in wide band
applications. What ferrites for HF
chokes? A HF choke will perform well up to the moment
of (parallel) resonance of the inductance with its parasitic parallel
capacity. Past this resonance frequency, this parasitic capacitance is
becoming the dominating impedance and this will fall with frequency up to the
moment total impedance is too low for the designed choking action. It
follows, that for wide band use, this parasitic capacitance should be as low
as possible, which usually translates into designing a choke with as low a
number of turns as possible. This in turn translates into selecting a ferrite
material with as high a (complex) permeability (mC) as possible over the operational frequency
range. In table 2 one may find various materials
exhibiting such high permeability over the HF range of frequencies, e.g. 3S4,
3F4, 4A11, 33, 43 4B1. The widely applied 4C65 (61) type is not in this list
since it is much easier to reach high impedance with less turns over this
frequency range with the lower frequency materials. In high(er) power systems though, one has to compare maximal
band-width performance (high m, less turns, low parasitics) to maximum core load (high Q, lower internal
core dissipation, lower temperature rise). We will come back to this in the
next chapter. For HF choke applications, ferrite type 4A11
(43) in general is the material of choice. Therefore one may also find this
material applied in the transformer in the Multiband trap antenna. From table 2 it may also be appreciated the higher
frequency materials 4C65, 61, and 68 to be more suitable for choke
applications above 30 MHz. because only here the
complex permeability tends to rise above the other materials. At these frequencies however
maximum load is also diminishing since materials loss factor is becoming
dominant above the ferrimagnetic resonance
frequency. How ferrites in HF chokes? We already noticed for high permeability
materials, every time a wire passes through the center hole in the toroide
coil former will account for a complete turn. Therefore, it is possible to
stack ferrites of different (ferrite) materials to design ultra wide
bandwidth chokes. The low frequency, high permeability material is dominant
at lower frequencies while the high frequency, lower permeability type takes
over at the high frequency side. This can have chokes perform at high and
constant impedance over very wide band width; this idea is taken somewhat
further in Ferrites in EMC applications. Sometimes one may come across a stack of an
orange core (3E25) and a purple core (4C65) for the purpose of achieving wide
band-width. In table 2 we find at 4 MHz., 4A11 (43)
type of ferrite to already exhibit a higher (complex) permeability than 3E25
type while this first material is also superior to 4C65 (61) at 30 MHz. It follows that a single 4A11 toroide choke with the
same number of turns is superior to the stack of the other two materials for
all of the HF amateur frequency bands. Stacking therefore usually only makes sense
for very wide bandwidth, including LF, MF and HF frequencies. It should be noted that stacking ferrite toroides is superior to constructing two chokes, each
with the same number of turns but on separate toroides.
At low frequencies no difference will be measured, but a high(er) frequencies one may be in for a surprise. Inductance
of one choke may resonate with the parasitic capacitance of the other,
creating a low impedance path across this combination instead of the high impedance
this system was designed for in the first place! Ferrites in HF transformers
Ferrites may be successfully employed in
various types of transformers, usually at higher that mains frequency. Ferrite cores may also be found in pulse
transformers (1- 100 kHz.), switch mode power supplies (20 - 1000 kHz.) and
HF voltage or impedance transformers in small- or wide-band applications (1 -
50 MHz.) In this chapter we will only discuss linear
wide band HF-transformers for small and medium size power applications. By
wide band we mean at least a few octaves in frequency and medium power up to
around 2 kW. More power is quite possible, but this will be outside this
story. For this chapter we will distinguish two main categories, based on
different principles: ·
Induction type of transformers based on magnetic field coupling for
energy transport. Core material is employed to concentrate the
electromagnetic field. This is the conventional type of transformer ·
Transmission-line transformers with energy transported by transmission
lines. Core material is employed for better input to output separation. Latter type will be discussed in two separate
chapters: Transmission-line
transformers, introduction and Transmission-line
transformers, examples and analysis. Basic induction type of transformer consists
of a primary and a secondary winding, usually around a magnetic core. This
core will concentrate magnetic field-lines to ensure both windings sharing
the same flux and so optimize transformer coupling. In this chapter
transformers will be constructed on ferrite type core material that mainly
determines transformer properties and application area. Next to inter-winding coupling, core material
will also enhance transformer inductance. Induction transformers are mainly
applied in parallel to the signal path and since the inductor current is out
of phase to the (resistive) load current, a phase shift will be introduced in
the signal path. We prefer this
disturbance to be as small as possible so the transformer inductance should
be high compared to the system impedance. To create a high impedance, we need a number
of turns around some ferrite material. This will inevitably create a
parasitic parallel capacitance across this transformer, that will further
increase with each next turn. Again, this capacitance will be parallel to the
transformer and so parallel to the signal path. To keep this influence small,
the number of turns should be kept small. To balance these contradicting effects we
like to apply high permeability material to keep inductance high while at the
same time keeping the number of turns low; this is where ferrite core
materials come in. Wide-band
transformers In figure 5 one will find the general lay-out
of a basic transformer, with turns ratio t = n2 / n1,
which is also the ratio of the voltages, or the reverse ratio of the
currents. The impedance ratio follows from t2.
An example of such voltage transformer for
wide-band HF frequencies is known as the 'Magnetic Longwire
Balun', MLB, that according to the commercials is recommended to match the
receiver to any long-wire antenna. This MLB actually is a 3 : 1
voltage transformer, usually constructed as an autotransformer for LW, MW and
SW frequencies. Looking into a few examples I
found most transformers to consist of a ferrite toroide 14 x 9 x The MLB is not constructed as a
balun (balance to unbalance transformer) but as an unun,
and not very 'friendly' to real long-wire impedances. This may not be a
problem since a long-wire antenna is at least one wavelength long with means
at least Stories may be told about all fantastic
performance claims around this component and also about precisely the
opposite. Whether friend or foe, all agree that this is a receiving-only
component, which makes us curious as to the performance in a transmitter
environment. As an example we will look at the performance
of the transformer in a 50 to 450 Ohm system as may be found at a Windom-type
of antenna at various HF bands. From the core dimensions we calculate core
volume (0,63 cm3) and from this maximum internal dissipation at 1 Watt (see Materials and
properties, Maximum core temperatures).
In table 7 we calculated primary impedance (Zp @ 10 turns) at various HF frequencies,
maximum allowable voltage for 1 Watt loss in the core (Udiss.,
formula 19) and maximum voltage for linear performance (Uinduction, formula 5).
Table 7. MLB, maximum voltages at 50 Ohm side We find the transformer already to exhibit
high impedance at the lowest HF frequency. In chapter 2 we found this
parallel impedance should be at least four times system impedance (200 Ohm in
a 50 Ohm environment) which will make this component's lower cut-off
frequency (200 / 1036) x 1,5 MHz.) = 290 kHz, which
is somewhere in the LW broadcast band. Third columns is showing that the transformer
may be applied at up to 44 volt across, which allows this component to be
applied at up to 38 Watt system power in a 50 Ohm system, or even at over 100
Watt PEP when only SSB type of modulation is employed. This ensures the MLB
may be safely employed at a small transmitter to well over QRP levels,
especially since maximum voltage for non-linear behavior is much higher at
minimum 70 volt (last column). We should keep in mind though that the maximum
voltage for internal dissipation is based on a component that is free to
radiate. Some of the MLB's however have been encapsulated in (isolating)
compound, lowering the internal dissipation. Please note since the MLB is no balun we will
have to add a good balun when applying this transformer at a Windom antenna
as in our example. The high impedance as in column two has been
obtained by the high permeability type of ferrite together with the high
number of turns (30 turns at the secondary side). At this small core, so many
turn will be very close together and even overlap, making parasitic parallel
capacity relatively high. Transformer behavior will therefore drop-off
quickly at higher HF frequencies. Since this HF behavior is very much
depending on the exact way these 30 turns are being applied, each device has
to be assessed separately. Wide-band
transformers, a closer look
When designing wide-band transformers it
often is more convenient to operate with parallel circuits. Therefore we may
recalculate series values we were discussing up to now into parallel values: μp'= μs'
(1 + 1/Q2) en μp'' = μs''
(1 + Q2) (27) From these formula's we may quickly
appreciate that for Q > 3, parallel induction (μp')
is roughly equal to series induction, while the parallel loss (resistor, μp'') quickly rises to high values. As an
example we will calculate a simplified model of a 1 : 1 transformer and its
effects on circuit parameters. The model may be found in figure 6, after
applying formula 27 to the series values.
The transformer on the right hand side is
considered to be 'perfect' in this model. Transformer inductance Lp and loss RFp
are in parallel to the input with the load resistor Rb
at the secondary side, to be re-calculated to the input side (with the
perfect 1 : 1 transformer, Rb = R'b). This simplified model is allowed at
lower HF frequencies, with high-permeability (ferrite) core material and not
too many turns ( low parasitic capacitance). Let's calculate this transformer when
constructed with two times 6 turns on a Behavior at 1,5 MHz In table 2 we will find for this material
that ĩ = 0, meaning no loss. Since 4C65 is a high frequency material, we may
use the factory value of XLp
= XL = j
w L = j w n ē This reactance is in parallel tot the load resistance Rb
(50 Ohm), when transformed to the left-hand side (same value, 1 : 1
transformer). Total input impedance: ZT = j.XL
. Rb / (j.XL + Rb)
= 28,6 + j.24,7 (note: complex
multiplication / division!) This is different from system impedance
without the transformer, when the 50 Ohm load was ensuring characteristic
termination and so SWR = 1. We have to re-calculate SWR with the new termination
value of ZT while being aware this to be a complex calculation
instead of the simple and better known formula SWR = Z0 / R (or
reverse). SWR = ( |Z0 + ZT| + |Z0
- ZT| ) / ( |Z0+ ZT| - |Z0 - ZT|)
= 2,32 (vertical bars express absolute values) If this is not acceptable, we have to enhance
the number of turns while keeping in mind that more turns also mean higher
parasitic capacitance so earlier fall-off at the high frequency side. With
the transformer as is (6 turns), SWR will be 1,5 at 3,5 MHz.,
which usually will be considered as the lower application limit. Behavior at 30 MHz. In table 2 we will find ĩ = 150 and ĩ = 45,
which means Q = 3,33 (low) and we will have to re-calculate basic values. First
we calculate the shape factor F using formula 8: F = m0 . A / l
= AL / mi , after which reactance and loss may be
calculated according to formula 11 and 12: XLs = j
w
L = j.w.nē.m
.F = j.2.p.30.106. 6ē.150.1,36.10-9 = j.1020 W RFs =
w.nē.m .F =
2.p.30.106. 6ē. 45.1,36.10-9 =
305 W Since these are series values, we recalculate
to parallel with formula 27: XLp = XLs (1 + 1/Qē) = j. 1380 (1 +
1/3,33ē) = j.1110 W RFp
= RFs
(1 + Qē) = 415
(1 + 3,33ē) = 3700
W With Rb
= 50 Ohm in parallel, total input impedance is 49,3 Ohm with negligible
inductance in parallel. This will make SWR = 1,05 which represents an almost
ideal situation if not for the parasitic parallel capacitance, we will leave
out for now. We might have considered constructing this
transformer on the same toroide size, but different ferrite material e.g.
4A11. At a frequency of 1,5 MHz. we would have
found parallel values XLp = j. 425 W and RFp
= 2250 W., (calculation as above, try for
yourself). This will yield SWR=1,13, meaning this transformer is already much
better at 1,5 MHz. than our earlier design on 4C65
material at 4 MHz. At a frequency of 30 MHz.
we find parallel values for 4A11 material
XLp = j.
9350 W and RFp =
2430 W., making SWR = 1,02, which makes
this transformer as good as the 4C65 design. This example is showing that different
ferrite materials will really make a difference when designing transformers.
Depending on the particular application, other system parameters have to be
taken into account as well. These basic calculation have been showing how to
start the design and how to apply table 2. Model
of an induction transformer In figure 6 we saw a first order model that
may be applied with satisfactory results at low(er)
frequencies. For a better model, also copper loss should be incorporated as a
series resistance with the transformer. Since HF transformers usually employ
a limited number of turns, this copper loss may usually be left out of the
model. Next non-ideal factor is the loss of magnetic coupling. At higher
frequencies and especially past the ferrimagnetic
resonance frequency, permeability is going down allowing flux-lines to escape
from the core center. This coupling loss may be measured at the desired
frequency by determining input inductance while shortening the output. Coupling loss may be incorporated in the
model as a series inductance, known as leakage inductance. The effect of this
series inductance may be measures as a high-frequency roll-off. Various schemes are going around to reduce HF
leakage: tight winding and / or twisting secondary and primary windings
before winding the transformer. All of these methods will indeed enhance
inter-winding coupling but will also enhance parasitic capacitance across the
windings which has the same high-frequency roll-off effect. A better way is to spread the primary winding
across the entire circumference of the toroide and placing the secondary winding
in between the primary turns, like a screw within a screw. Best to start with
the winding with the lowest number of turns. When the transformer is ready,
windings are spread-out evenly at the inside of the toroide tot minimize parasitic capacitance. The effects of this
winding technique will be already noticeable at 15 MHz.
and more so at higher frequencies. The better model should also contain a
capacitance across both inductances. The effect of this parasitic capacitor
is almost negligible at lower frequencies
(< 10 MHz, at Z < 100 W), but has to be taken into
account at higher frequencies as this usually is one of the important
parameters for high-frequency roll-off. Capacitor values may be determined
from parallel resonance of primary and secondary winding. In figure 7, non-ideal effects as discussed
have been incorporated in the transformer model.
Symbol table: Rb: Load impedance, transform to primary by
division by n2 Cs:
Transformed secondary winding parasitic capacity Lp:
Parallel inductance (from unloaded transformer measurement) RFP: Total core loss Ls:
Leakage inductance Rs:
Generator output impedance U:
Generator From this model it is obvious that a
spread-sheet program should be applied when calculating transformer behavior
over frequency. Please keep in mind this model is no different from LF transformer
models as applied to microphone transformers, LF tube / transistor input- and
output-transformers, modulation transformers etc. A special type of transformer arises when
primary turns and currents are equal to secondary turns and currents.
Principle is modeled in figure 8. .
In this induction transformer, currents i1
and i2 are equal and in opposite phase. With also the number of
turns to be equal, the magnetic fields in the transformer core will cancel.
This effect will be found at bifilarly wound transformers and also at
transformers constructed around transmission-lines, either balanced or
coaxial. The current balancing effect is fully
comparable to what is happening inside a transmission-line. This current
transformer therefore is also known as a transmission-line transformer. Basic
principles are the currents to be equal and in opposite phase and so the
voltage of the generator at the input will also be found at the output of the
transformer and across the load. Voltage induction-transformer in general are
lagging behind to current transformers as far as power transfer is concerned.
Since a voltage transformer is always in parallel to the signal path, a
(small) portion of power is dissipated inside the transformer. Furthermore,
at un-equal turns ratio's, internal transformer fields do not cancel any more
making this type of transformer prone to non-linear effects at high voltage
and power load. In current transformers, parasitic core
currents may only originate from un-balancing effects, which we try to make
as small as possible by design and / or by minimizing outside return currents
by means of high common-mode impedance. A more elaborate description will be found in
the next chapter on Transmission-line
transformers, introduction. Bob J. van Donselaar, on9cvd@veron.nl |
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